Advanced data encoding and modulation techniques have made it possible to transmit ever increasing amounts of information through radio frequency (RF) wireless communication systems. Accurate power measurement and control is critical to implementing these schemes. For example, code division multiple access (CDMA) systems rely on a coding scheme in which multiple users occupy the same part of the frequency spectrum, but to one user, signals from other users appear as noise. For such a system to work effectively, the transmitted power levels from individual handsets must be controlled so that the transmitted signals are received at the base station at about the same power level, regardless of how far each handset is from the base station. CDMA signals also have relatively high peak-to-average power ratios. These characteristics place extreme demands on the power measurement and control system.
Another transmission technique known as orthogonal frequency division multiplexing (OFDM) relies on multiple orthogonal subcarriers. Each subcarrier has a relatively low amplitude, but since numerous subcarriers occasionally add in phase to create very high instantaneous amplitudes, the resulting OFDM waveforms have very high peak-to-average power ratios.
Systems utilizing modulation schemes with high peak-to-average power ratios must often be operated in a condition known as “backoff” in which the output of the power amplifier is reduced to prevent distortion that occurs when the signal peaks exceed the linear range of the amplifier. Accurate power measurement is critical for implementing backoff control schemes.
Several different types of detectors are commonly used to measure the power of RF signals. They range from simple diode-detectors to more complex logarithmic amplifiers (log amps) and root-mean-square (RMS) detectors. The measurement characteristics of these detectors typically vary with temperature. For relatively low operating frequencies, the temperature dependencies of a detector can usually be corrected by a compensation signal having a relatively simple temperature function. As the operating frequency increases, however, the simple temperature functions become inadequate, and more complicated adjustments are required.
FIG. 1 illustrates a prior art temperature compensation circuit for an RF power detector. Currents I3 and I4 create voltage drops across the base-emitter junctions of diode-connected transistors Q3 and Q4. The difference between the base emitter voltages (ΔVBE) is applied as the input voltage Vi to a transconductance multiplier cell formed by Q1 and Q2. The ΔVBE is multiplied by the tail current I5 to produce a compensation signal IOUT which is taken as the difference between currents I1 and I2.
In one configuration, the circuit of FIG. 1 is arranged in combination with the circuit of FIG. 2 to provide a compensation signal having a logarithmic temperature function that can be used to temperature correct the output of a log amp. In this configuration, the current I3 is made proportional to absolute temperature (PTAT), while I4 and I5 are implemented as temperature-stable currents (also referred to as ZTAT currents where the Z represents zero temperature coefficient). This particular combination of PTAT and ZTAT currents for I3 and I4 creates a ΔVBE having the form VT ln(IP/IZ). When this ΔVBE is applied to Q1 and Q2, the resulting output has the form IOUT=I5 tan h[ln(IP/IZ)] which can be approximated as IOUT≈I5 ln(IP/IZ).
The circuit of FIG. 2 includes a user-accessible terminal ADJ to enable the user to vary the amount of tail current I5, and thereby adjust the amount of temperature compensation based on the operating frequency. The reference voltage VREF is a temperature stable reference voltage, so the current through R3 is also temperature stable (ZTAT) and varies only with the value of RADJ which the user connects to the terminal ADJ. Therefore, the tail current I5 applied to the circuit of FIG. 1 varies only with the value of RADJ. The manufacturer typically provides a table of suggested values for RADJ for various common operating frequencies. This type of temperature compensation scheme is further described in U.S. Pat. No. 7,180,359 which is by the same inventor as the present patent disclosure and is incorporated by reference.
In another configuration, the circuit of FIG. 1 is used in combination with the circuit of FIG. 3 and arranged to provide different types of temperature compensation as might be suitable for other types of detectors such as RMS detectors. For example, if I4 is made complementary to absolute temperature (CTAT) rather than ZTAT, then the curvature of the ΔVBE function is more pronounced than in the example above and provides a greater amount of adjustment. Additional shaping of the temperature function can be provided based on the form of the tail current I5. For example, if I5 is implemented as a PTAT current, the curvature of the ΔVBE function is even more pronounced. Alternatively, however, the tail current I5 can be made CTAT, ZTAT, or any other function of temperature.
As with the circuit of FIG. 2, the circuit of FIG. 3 also includes a user-accessible terminal ADJ to enable the user to vary the amount of tail current I5, and thereby adjust the amount of temperature compensation based on the operating frequency. In this case, however, the temperature shape of the tail current I5 is determined by the temperature shape of I6. For example, if I6 is PTAT, the tail current I5 is also PTAT. By varying the voltage applied to ADJ terminal relative to the reference voltage VREF/2, the amount of tail current applied to the multiplier can be adjusted to provide an approximation of the required temperature compensation for the specified operating frequency.
FIG. 4 illustrates another prior art circuit for providing temperature compensation to an RMS detector. The temperature dependency is provided by the junctions of Q8 and Q9. The currents through Q8 and Q9 pass through cascode transistors Q10 and Q11 and are then mirrored by current mirrors Q12,Q13 and Q14,Q15 to another mirror Q16,Q17. The difference current ICOMP is used to affect the bias of the detector, for example by applying ICOMP to a resistor ladder that is used in an interpolator for an RMS detector having a multiplicity of squaring cells. The user can adjust the compensation by connecting resistors Rcpa and Rcpb to user-accessible terminals CPA and CPB. Any difference between the currents Icpa and Icpb causes a negative or positive tilt in the temperature compensation. Trial-and-error techniques must be used to determine values of Rcpa and Rcpb that provide an approximation of the amount of temperature compensation required for any particular operating frequency.